The present invention generally relates to DCxe2x80x94DC converters and particularly, to an improved circuit topology for boost converters.
A boost converter is a type of flyback converter where a smaller input DC voltage is increased to a desired level. A prior art typical boost converter 10 is shown in FIG. 1. As shown in FIG. 1, the typical boost converter includes an inductor 15, to which the input voltage Vin is coupled, that is in series with a boost diode 16 connected to an output capacitor 22 across which the load 19 is connected. A transistor switch 14 is connected to a node 12 between the inductor 15 and boost diode 16 and ground to provide regulation of the output voltage. The control circuit 20 for the transistor switch 14 typically includes a comparator (not shown) for sensing and comparing the output voltage of the converter to a voltage reference to generate an error voltage. This error voltage is then coupled to a duty cycle constant frequency pulse width modulator circuit (PWM). The PWM converts the error voltage into a control signal. A gate resistor 18 couples this control signal to the transistor switch 14 control input for controlling the timing of the on and off transition of the transistor switch 14. When the transistor switch 14 is on, the inductor current increases, storing energy in its magnetic field. When the transistor switch 14 is off, energy is transferred via the diode 16 to the output energy storage capacitor 22 and the load 19. The transistor switch 14 is operated at a high frequency relative to the resonance of the inductor capacitor network.
Drawbacks of such conventional boost converter circuits include the creation of switch voltage and current stresses resulting in low efficiency power conversion. Another drawback of switched power circuits is the electromagnetic interference (EMI) arising from the large change in current (di/dt) and voltage (dV/dt) that occurs when the switch changes state. More specifically, one drawback of the conventional boost converter circuit in FIG. 1 is the recovery current of the boost diode 16 added to the power loss due to the discharge of the switch output capacitance, Coss, of the switch 14 (1/2 Coss V2f) at turn on. Increased EMI noise is also generated due to the snap off of the boost diode 16 after it stops conducting. Another drawback of the boost converter 10 is the losses at turn off of transistor switch 14. The Coss of the transistor switch 14 is so low that the turn off loss is significant. Increasing this capacitor does not overcome this because the losses would only be transferred at turn on. To overcome these drawback, boost converters have been proposed that provide soft switching, i.e., switching at low voltage and current stress across the transistor switch. A prior art boost topology 30 to overcome the drawback at turn on of the main switch 14 is shown in FIG. 2.
As shown in FIG. 2, an input voltage VIN is converted into output power (VOUT) using a resonant network in addition to the conventional components of a boost converter. The resonant network comprises a snubber inductor 32, coupled in series with resonant diode 34 and a 36. Auxiliary switch 38 and resonant diode 36 are in series and are connected in parallel with main switch 14. The snubber inductor 32, with a value significantly smaller than the boost inductor 15, in conjunction with the auxiliary switch 38 is added to control the recovery current of the boost diode 16 at its turn off. This topology allows a zero voltage switching (ZVS) on the main switch 14 and a zero current switching (ZCS) on the auxiliary switch 38. In operation, a ZVS detection circuit (included in control circuit 44, details not shown) monitors the voltage across the main switch 14 to turn it on at zero volts. Snubber inductor 32 limits the current at turn on of the auxiliary switch 38 to achieve the ZCS.
A drawback exhibited by the boost topology of FIG. 2 is that the energy in the parasitic capacitor of resonant diode 34 is transferred to the snubber inductor 32 at the turn off of the main switch 14. This transfer results in a current in resonant diode 34 that turns resonant diode 36 on. When the auxiliary switch 38 is turned on, the recovery current of resonant diode 36 will generate a current spike that causes losses in the auxiliary switch 38 and increased EMI noise. The power lost through the auxiliary switch 38 reduces the efficiency of the boost converter. A boost topology 40 to overcome the drawback associated with turn off of the main switch 14 and auxiliary switch 38 is shown in FIG. 3.
The boost topology 40 in FIG. 3 adds a snubber capacitor 42 and resonant diode 44 to the topology shown in FIG. 2. In operation, at turn off of the main switch 14, the snubber capacitor 42 is already charged to the output voltage. As a result, the current circulates into snubber capacitor 42 and the resonant diode 44 to smooth the dv/dt across the main switch 14. The snubber capacitor 42 will discharge to zero in order to turn the boost diode 16 on. At turn off of the auxiliary switch 38, the series combination of the discharged snubber capacitor, resonant diode 36 and main switch 14 are in parallel with the auxiliary switch 38 and smooth the dv/dt. The snubber capacitor 42 will again be charged to the output voltage. For this operation, energy is only exchanged between the Coss of each switch via the snubber capacitor 42, thus there is no additional energy dissipation. The topology of FIG. 3 addresses the dv/dt at turn off the switches, however, a drawback exhibited by this topology is associated with losses due to the recovery current of resonant diodes 36 and 44 at turn ON of the auxiliary switch. A boost topology 50 to overcome this drawback is shown in FIG. 4.
As shown in FIG. 4, a boost topology 50 adds an inductor bead 52 and a clamping circuit formed by diodes 54 and 56 to the topology of FIG. 3. A commonly assigned U.S. Pat. No. 6,236,191 ZERO VOLTAGE SWITCHING BOOST TOPOLOGY which is incorporated by reference herein. U.S. Pat. No. 6,236,191 discloses a topology similar to boost topology 50 without the clamping circuit. This topology adds the inductor bead 52 in conjunction with the slower resonant diodes 36 and 44, to overcome the drawback of recovery current of those diodes at turn on of the auxiliary switch 38. For this topology, resonant diode 34 is a fast recovery type diode, such that it stops conducting (and recovers) before the remaining resonant diodes. The remaining charges in the slower resonant diodes 36 and 44 begin to charge the parasitic capacitor of resonant diode 34. In operation, resonant diodes 36 and 44 must be slow enough to ensure that resonant diode 34 recovers first, but fast enough to be recovered before the parasitic capacitor of resonance diode 34 is charged. If resonant diodes 36 and 44 are not recovered when the parasitic capacitor of resonant diode 34 is fully charged, a current spike will occur upon turn ON of the auxiliary switch 38. This charge up of the parasitic capacitor of resonant diode 34 is completed by the parasitic capacitor, Coss, of the auxiliary switch 38. However, because resonant diodes 36 and 44 are slower than resonant diode 34, the parasitic capacitor, Coss, of the auxiliary switch 38 will not discharge as much as if the diodes were the same. This reduces the resonance between the snubber inductor 32 and Coss of the auxiliary switch which reduces the current in resonant diodes 36 and 44.
As the first resonant diode 34 is a fast recovery type diode, it recovers the stored charge that is dissipated by the snubber inductor 32 and stops conducting the corresponding current before the second and third resonant diodes 36,44 recover their stored charges. In this fashion, the current flowing through the first resonant diode 34 and into auxiliary switch 38 when it turns ON during its next cycle is substantially eliminated. As a result, power losses associated with the auxiliary switch 38 turning ON are substantially eliminated, as well as EMI noise reduced.
In the topology in FIG. 4, a clamping circuit is formed by the series combination of clamping diode 54 and zener diode 56. This combination is coupled between the junction of snubber inductor 32 and first resonant diode 34 and ground. In operation, clamping diode 54 and zener diode 56 clamp the voltage to prevent first resonant diode 34 from reaching its breakdown voltage in a high ambient environment due to the increase of recovery current and the saturation of the bead 52.
A drawback exhibited by the topology of FIG. 4 is that its effectiveness is affected under high ambient temperature due to the large variations (deltas) in the BH curves from one bead manufacturer to another. Because of this variation, in some cases, the second and third resonant diodes 36,44 are still conducting when the auxiliary switch 38 is turned ON, resulting in a current spike, increased power loss and EMI. An additional drawback of the topology of FIG. 4 is the speed requirements for the resonant diodes. Second and third resonant diodes 36,44 must be slower compared to first resonant diode 34, but not too much slower. They must be fast enough to be recovered before the parasitic capacitor of resonance diode 34 is charged. Another drawback of the topology of FIG. 4 is that the second and third resonant diodes 36,44 must have the same temperature behavior characteristics in terms of rapidity in order to stay in the same ratio as compared to the first resonant diode 34.
The aforementioned and related drawbacks associated with prior art boost converters are substantially reduced or eliminated by the improved boost converter topology of the present invention.
The present invention improves upon the topology in FIG. 4 by adding a secondary winding to the boost inductor. This winding is connected in series with a resonant diode that connects to the drain of the auxiliary switch. A capacitor has also been added in parallel with a resonant diode connected in series with the secondary winding. The present invention has the advantage of reducing the energy stored in the parasitic capacitor of the first resonant diode by a factor of 4 at the turn off of the main control switch. This reduction is achieved by allowing only a small amount of energy transfer to the boost snubber inductor so that it does not turn the second and third resonant diodes on before the auxiliary switch is turned on, thus reducing losses and EMI.
For the present invention, at turn off of the main control switch, the voltage is shared between the two parasitic capacitors of a resonant diodes connected to the primary and secondary windings of the boost snubber inductor. During this time, no energy is stored in the boost snubber inductor due to the transformer effect. The capacitor added in parallel with the resonant diode connected to the secondary winding will discharge even more of the parasitic capacitor of the diode connected to the primary winding.
In a preferred embodiment of the present invention, the boost converter comprises a power converter having two input terminals for connection to a power source, comprising an inductor connected to a first one of the input terminals; a boost diode connected in series communication with the inductor, the boost diode having a cathode connected to a first output terminal; an output capacitor coupled across the first output terminal and a second output terminal; a control switch, connected between a first node at the junction of the series connected inductor and boost diode and a second of the input terminals, for controlling the application of voltage to the output terminals as a function of the duty cycle of the control switch, the second of the input terminals and the second output terminal being connected in common; a resonant network, coupled between the first node and the first output terminal, the resonant network including a snubber inductive element, a first resonant diode, a second resonant diode, a third resonant diode, and a fourth resonant diode, the snubber inductive element having a primary winding and secondary winding, the primary winding and the first resonant diode connected in series between the first node and a second node; the cathode of the first resonant diode coupled to the second node and the anode of the first resonant diode coupled to the primary winding; the second and third resonant diodes connected in series between the second node and the first output terminal, the anode of the second resonant diode coupled to the second node and the cathode of the second resonant diode coupled to the anode of the third resonant diode, the secondary winding and the fourth resonant diode connected in series between the second input terminal and the second node; the anode of the fourth resonant diode coupled to the secondary winding and the cathode of the fourth resonant diode coupled to the second node; a first resonant capacitor coupled between the first node and the junction of the second and third resonant diodes; and an auxiliary switch, connected between the second node and the second output terminal, the auxiliary switch being turned on such that the voltage at the control switch is caused to reduce to zero when the control switch turns on.
The improved topology of the present invention thus allows for the control switches to be turned on without exhibiting the level of power losses experienced by prior art boost converters.
An advantage of the present invention is that it provides the ability to turn on the control switches of a power converter without producing significant power losses.
Another advantage of the present invention is that it provides for zero voltage switching of the control switches of a power converter.
Yet another advantage of the present invention is that it reduces the amount of stress exhibited with respect to the components of a power converter and the resultant EMI noise.
Another advantage of the present invention is the effectiveness of the topology is not affected under high ambient temperatures due to the variation in BH curves of saturable beads among manufacturers.
Another advantage is that the topology of the present invention allows the use of the same diode type. Thus the present invention does not depend upon the precise selection of resonant diode speed characteristics, and is not dependent upon those speed characteristics being consistent over the operating temperature ranges.